Methods and apparatus for providing reciprocal impedance conversion

ABSTRACT

A reciprocal impedance conversion network is disclosed. Such conversion network preferably is used in a reciprocal negative impedance repeater for the nonloaded cable facilities of a telephone system. Two independent voltage sources for generating reciprocal negative impedance are connected between a first port and a second port. The first port is operably connected to the first voltage source such that a series negative impedance appears at that port. This port is specifically connected to the switching equipment of the telephone system. The second port is operably connected to the second voltage source such that a shunt negative impedance appears at that port. The second port is specifically connected to the nonloaded cable facilities of the telephone system. The invention further provides a frequency dependent gain circuit which is useful for equalizing the losses along the nonloaded cable facilities. In addition, the invention provides an unconditional stability test for the reciprocal negative impedance repeater which may be performed substantially without test equipment.

This is a continuation, of application Ser. No. 117,251, filed Nov. 4,1987.

The invention relates generally to methods and apparatus for reciprocalimpedance conversion, and is more particularly directed to such methodsand apparatus used in reciprocal negative impedance repeaters adaptedfor use with nonloaded cable facilities of a telephone system.

Because of losses in many instances it is necessary for the cablefacilities of a telephone system to include the amplification of aninformation signal. Amplification of subscriber signals is accomplishedin the voice frequency band by a device usually referred to as a voicefrequency repeater, VFR. These devices are also termed more broadly2W--2W (two-wire) repeaters. Previously, amplification of voicefrequencies in two-wire telephone systems has been achieved using the ofthe hybrid transformer repeater or by negative impedance techniques. Toachieve the high degree of cable impedance matching necessary for stableoperation of either repeater type, it is generally required that amultitude of precision balance networks or line build out sections beused. The cost and complexity of these sections, in addition to theirassociated alignment problems is much greater than is necessary.

In characterizing negative impedance repeaters, the prior art includesseries and shunt negative impedance elements as well as combinationsthereof. A widely adopted configuration is the series-shunt bridged-Tarrangement which is adapted to provide a relatively fixed overall gainwhile maintaining a predetermined image impedance. This negativeimpedance repeater utilizes a passive impedance matching network tocorrect the line impedance which operates in conjunction with abridged-T negative impedance gain unit. The impedance matching networkis not only adapted to match impedance but also attenuates lowfrequencies by approximately the same amount that high frequencies areattenuated by the cable facilities. This produces an amplitudeequalization, but the combined attenuation of the line and the impedancematching network is relatively substantial such that large portions ofthe available repeater gain are used only to overcome these losses.

An alternative approach provides a bidirectional bridged-T negativeimpedance repeater having provisions for matching the image impedance ofthe repeater to the characteristic impedance of the cable. Such systemsare difficult to install and align as different gain and impedancesettings are required for every length and gauge of cable. Additionally,as the image impedance of the repeater is adjusted to match theimpedance of the cable facilities, a maximum power transfer cannot beobtained in normal repeater applications where one port of the repeateris directly coupled to terminal switching equipment having acharacteristic impedance of approximately 600-900 Ohms in series with2.15 microfarads.

Alignment of the bridged-T negative impedance repeater is furthercomplicated because of its combination of series-shunt negativeimpedance elements. To perform an unconditional stability test on a twoport repeater both ports must be tested with "worst case" conditions.The worst case condition for a series negative impedance is a shortcircuit and the worst case condition for a shunt negative impedance isthe opposite, an open circuit.

However, because bridged-T negative impedance repeaters exhibit bothtypes of impedances at both ports, a difficult four step unconditionalstability test is necessitated. All combinations, short on the firstport-short on the cable facilities, short on the first port-open on thecable facilities, open on the first port-short on the cable facilities,and open on the first port-open on the cable facilities must beattempted before the repeater can be certified as unconditionallystable. The short circuiting of the cable facilities is particularlydifficult because it requires a test person to actually be present atthe end of the cable loop to execute the test. Because of thesensitivity of this type of repeater such time consuming alignments maybe required for even small changes in network configuration.

SUMMARY OF THE INVENTION

The invention provides a reciprocal impedance conversion network havinga first port and a second port for bidirectional transmission through animpedance conversion means. The network is configured such that a signalgenerator, having a source impedance connected to one port, is able totransmit to a load impedance at the other port. The impedance conversionmeans connected between the first and second ports provides an inputimpedance at the transmitting port which is inversely proportional tothe load impedance at the receiving port. The load impedance can bedifferent combinations of negative or positive impedances in series orshunt with the receiving port which are then reciprocally reflected tothe transmitting port.

In one preferred embodiment the impedance conversion means comprise apair of voltage sources including a first voltage source for generatinga first control voltage and a second voltage source for generating asecond control voltage out of phase with the first. The first controlvoltage is generated proportionally to a first control impedance and thecurrent output by the source generator. This first control voltage isapplied to the receiving port and drives the load. The second controlvoltage is generated proportionally to a second control impedance andthe current drawn by the load impedance. The second control voltage isinverted to be of the opposite polarity of the first control voltage andis applied to the transmitting port. The first control voltage drivesthe load impedance proportionally to the current generated from thesource and the second control voltage measures the current through theload and provides a voltage to the first port which is inverselyproportional to the load impedance. For transmission in the oppositedirection, the roles of the voltage sources reverse.

According to the invention, another preferred embodiment providesnegative feedback of a fraction of the first control voltage incombination with the second control voltage to provide a series negativeimpedance at the first port which is inversely proportional to the loadimpedance. The second control voltage, because of this feedback, appearsas an independent shunt negative impedance which is inverselyproportional to the load impedance at the second port. In thisconfiguration the impedance conversion network provides gain andlinearity in either transmission direction.

This embodiment is particularly well adapted for use as a reciprocalnegative impedance repeater for nonloaded cable facilities of atelephone system. In this embodiment, the port exhibiting the seriesnegative impedance is connected to the switching equipment of atelephone system and the port exhibiting the shunt negative impedance isconnected to the nonloaded cable facilities of the telephone system. Theswitching equipment port is open circuit stable and short circuitunstable while the output cable port is short circuit stable and opencircuit unstable. Because the two voltage sources are substantiallyindependent, the unconditional stability of this repeater may be testedby the expedient of a single test where the cable facility is opencircuited and the switching equipment port is shorted. Additionally,because the cable port is short circuit stable and the cable impedancefor nonloaded cable decreases with additional subscribers, the changingof the cable facilities does not adversely affect the stability of therepeater. Moreover, a shunt across the cable facility, such as by thehandset of a lineman, will not adversely affect the stability of therepeater.

According to the invention, a virtual unconditional stability test canbe performed for a repeater of this type by merely shorting theswitching equipment side of the repeater at a time when the far end ofthe cable facilities are open circuited. The test takes advantage of thefortuitous fact that it is a normal condition for nonloaded cable to beopen circuited at the far end. This is the condition when the subscriberline is idle (subscribers "on hook" during nonuse or by request, orbecause none are connected to a new cable, etc.) Therefore, there is noneed to physically make any changes to the cable port side of therepeater to make this test and consequently no need to have testpersonnel present, other than at the repeater site.

The invention further provides a frequency dependent gain adjustmentmeans by which gain can be increased or decreased in accordance withcable gauge or cable length. A method of electronic alignment for theinvention is provided by this gain adjustment whereby during theunconditional stability test, i.e. while the switching equipment port isshorted, a visual indication from a singing detector is used todetermine whether the repeater is oscillating. The gain is increasedincrementally to where the repeater begins to sing. In a preferred formthe gain control includes a slide switch which incrementally addsresistance. When during the electronic alignment the repeater beginsoscillating, the slide switch is backed off one increment such that theoscillation ceases. In this manner the repeater is set for stability atits "worst case" and all other changes cause stability to increase.

Accordingly, it is a major object of the invention to provide aimpedance conversion network including the capability of providingbidirectional reciprocal impedance.

It is further an object of the invention to provide a impedanceconversion network useful in transmission equipment designs,particularly for reciprocal negative impedance repeaters in the voicefrequency range adapted for use with nonloaded cable facilities of atelephone system.

It is further an object of the invention to provide a reciprocalnegative impedance repeater which is electronically alignable forgain-equalization settings substantially without the use of testequipment.

Still further is an object of the invention to provide a reciprocalnegative impedance repeater for use in the voice frequency band withnonloaded cable facilities with improved stability criteria.

Yet another object of the invention is to provide a reciprocal negativeimpedance repeater which is linear in operation.

Still another object of the invention is to provide a reciprocalnegative impedance repeater which does not need impedance matchingnetworks or line buildout networks.

These and other objects, features and aspects of the invention will bemore fully understood and better described if a reading of the followingdetailed description is undertaken in conjunction with the appendeddrawings wherein:

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is an electrical system block diagram of a network havingreciprocal impedance generation means constructed in accordance with theinvention;

FIGS. 1A-1D are further embodiments of the network illustrated in FIG. 1for generating reciprocal negative impedance.

FIG. 2 is an electrical schematic block diagram of a reciprocal negativeimpedance repeater adapted particularly for use with nonloaded cablefacilities of a telephone system;

FIGS. 3A-3D are electrical schematic views of preferred implementationsof circuitry used to embody the impedance conversion means illustratedin FIGS. 1 and 2;

FIG. 4 is a detailed electrical schematic diagram of the reciprocalnegative impedance repeater illustrated in FIG. 2 implemented by theimpedance conversion means illustrated in FIGS. 3B and 3D;

FIG. 5 is a detailed electrical schematic diagram of the reciprocalnegative impedance repeater illustrated in FIG. 4 illustrating furtherrefinements for stability and frequency dependent gain equalization; and

FIG. 6 is a graphical representation of loss as a function of frequencyfor a nonloaded cable facility compared with a repeater conditionedsignal in accordance with the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

In the art of transmission lines, a negative impedance inserted inseries with a source and load impedance serves to decrease the impedanceof the load presented to the source by increasing the source current.The negative impedance appears to provide a gain which is directlyproportional to the magnitude of the negative impedance. As a conditionof stability the magnitude of the negative series impedance must alwaysbe less than the magnitude of the sum of the load and source impedance.Violation of this condition causes the system to become unstable and tooscillate.

Conversely, a negative impedance inserted in shunt in a network servesto increase the load voltage and thereby increases the apparent loadimpedance presented to the source. Accordingly, the gain provided by ashunt negative impedance is inversely proportional to the magnitude ofthe negative impedance. As a condition of stability, the magnitude ofthe negative shunt impedance must always be greater than the parallelcombination of the source and load impedance. Violation of thiscondition causes the system to become unstable and to oscillate.

It is evident that the application of either series or shunt negativeimpedance to transmission lines to provide gain without oscillation isextremely dependent upon the predictability of the transmission lineimpedance characteristics, particularly as those characteristics applyto the stability criteria.

Inductively loaded, or simply loaded telephone transmission cable isimplemented by introducing lumped inductance into the transmission lineat intervals to maintain a characteristic line impedance at a nominal600-900 ohms in series with 2.15 microfarads of capacitance. Thischaracteristic impedance is maintained across the audio frequency band,between approximately 300 and 3,000 Hz., and is substantiallyindependent of the length or gauge of the cable. The impedance ofnonloaded cable facilities is a function of the gauge of the cable aswell as the frequency of the signal applied to the cable by way ofcontrast..

As a general rule the attenuation, or loss, of telephone cable isdependent upon both the length and the gauge of the cable. In the caseof a loaded cable, however, the attenuation for a given length and gaugeremains nearly constant over the voice frequency band. This is in factthe reason why telephone cable is loaded. The attenuation characteristicof a nonloaded cable is frequency dependent attenuating higherfrequencies to a greater extent than lower frequencies.

Therefore, it will be seen for loaded transmission facilities that theimpedance and attenuation are both known and are both substantiallyfrequency invariant. This suggests a determinable amount of negativeimpedance (a repeater) may be inserted in the line with a reasonableexpectation that the system will remain stable. However, in the case ofnonloaded lines the frequency dependent characteristics of the cableintroduce appreciable difficulties in maintaining stability. Moreover,the nonloaded cable can appreciably change impedance when a subscribergoes from on hook to off hook.

Thus, the variation in impedance and loss which occur on nonloaded cablebecause of different cable size, different cable lengths, changes infrequency, and conditions such as subscriber calls or a lineman'shandset, all tend to destabilize a normal repeater. The invention, inone embodiment to be more fully described hereinafter, provides arepeater which is stabilized for the "worst case" of the variables andbecomes more stable as the conditions change.

One embodiment of the invention is shown to advantage in FIG. 1 where animpedance conversion network for bidirectional transmissions is shown.The conversion network 10 provides reciprocal impedance conversion fortransmission in either direction. The impedance conversion network 10includes a first Port A having terminals 12 and 14 which are connectedrespectively to the terminals of the primary of a one to one transformer16 whose secondary terminals 24 and 26 couple power to the network. Theimpedance conversion network 10 further includes a second Port B havinginput terminals 20 and 22 which are respectively connected to theprimary of a one to one transformer 18 whose secondary terminals 28 and30 couple power to the network.

A reciprocal impedance converter 33 is connected between thetransformers 16, and 18 and comprises impedance conversion means 32 and34. Impedance conversion means 32 and 34 generate independent reciprocalimpedances for each port. The impedance conversion means 32 generates avoltage Vab which is applied to the secondary terminal 28 of transformer18. The voltage Vab is generated by the impedance conversion means 32 asa function of the current I₁ in the secondary of transformer 16 and acontrol impedance Za. The preferred function in this embodiment is alinear function where:

    Za=Vab /I.sub.1

Impedance conversion means 34 generates a voltage Vba which is appliedto secondary terminal 24 of transformer 16. Voltage Vba is generated asa function of the current I₂ in the secondary of the transformer 18 anda control impedance Zb. Preferably, the function is a linear function ofthe form:

    Zb=-Vba/I.sub.2

In defining transmission direction, the following convention will beused. If a signal generator Esa having a source impedance Zlsa isapplied to Port A and a signal generator Esb is set equal to 0, thenimpedance Zlsb becomes a terminating or load impedance. These are thecharacteristics of a transmission in a direction from Port A to Port B.For transmission in the other direction, the signal generator Esb havinga source impedance Zlsa is applied to terminals 20 and 22 of Port B,where in this instance the generator Esa is set equal to 0 and Zlsabecomes the output or load impedance. These are the characteristics fora transmission from Port B to Port A.

Now, for transmission from Port A to Port B and assuming idealtransformers, the input impedance of Zin of Port A is:

    Zin=Za×Zb/Zlsb

Similarly the input impedance of Port B, assuming ideal transformers,is:

    Zin=Za×Zb/Zlsa

From the above equations it is shown that this linear network isbilateral when Za=Zb and the input impedance for each port is inverselyproportional to the destination or load impedance for each direction oftransmission. It is also evident from this analysis that the inputimpedances Zin for Port A and Port B can be scaled by the product Za×Zb.

When considering transmission from Port A to Port B the voltage Vbaopposes the flow of current I₁ in the secondary of transformer 16 andtherefore, provides a positive impedance. In this manner voltage Vbaappears as a positive impedance to the input of Port A. The voltage Vabon the other hand generates or causes current -I₂ and acts as a voltagesource to the output impedance Zlsb. In reversing the transmissiondirection such that information is transmitted from Port B to Port A,the control voltage Vab now opposes the current I₂ providing a seriespositive impedance and voltage Vba generates a negative current -I₁ todrive Zlsa. For the impedance conversion network illustrated in FIG. 1it is a unique feature that the voltages Vab and Vba toggle betweenbeing an impedance (current opposer) and a source (current producer) asthe direction of the transmission changes. For simultaneoustransmissions in both directions, of course, they operate to performboth functions.

If we consider transmission from Port A to Port B with generator Esb setequal to 0, then the output current I₂ is:

    -I.sub.2 =(Esa×Za)/(ZaZb+Zlsa Zlsb)

Likewise for transmission from Port B to Port A, with generator Esa setequal to 0, the output current I₁ is:

    I.sub.1 =(Esb×Zb)/(ZaZb+Zlsa Zlsb)

The equations illustrate that if Za is not equal to Zb, then the circuitis not bilateral and gain (or loss) are toggled by the scaling of Zawith respect to Zb. Furthermore, if the product Za×Zb is kept constantthe toggling of gain and loss can be done without changing the inputimpedance to transmission. When the voltage Vab is a voltage source itis in phase with the voltage Vba, and when the voltage Vba is a voltagesource it is out of phase with voltage Vab. The implementation shown inFIG. 1, when Za and Zb are not equal, can be used to implement aswitched gain repeater where gain in one direction during transmissionis enhanced while loss in the other direction is increased, and viceversa.

In FIG. 1 it is seen that the reciprocal impedances are positive, i.e.,there will be an attenuation of the signals provided from the signalgenerators for transmission in either direction. For repeaters and othernetwork devices it is advantageous to have the input impedance of atransmission apparatus exhibit negative impedance or gain. The impedanceconversion means 33 is able to accomplish this task in a bidirectionalmanner while still providing the function of controlling the inputimpedance as inversely proportional to the load impedance.

FIGS. 1A-1D illustrate specific embodiments of the impedance conversionmeans 33 used in an impedance conversion network which exhibitsreciprocal negative impedance. Reciprocal negative impedance is anegative input impedance which is inversely proportional to the loadimpedance for the particular direction of transmission. FIG. IAillustrates a network having a Port A and a Port B where an impedanceconversion means 33 is connected between the two ports. A negativeimpedance -Z1 is connected in series with Port A. The impedanceconversion means 33 inverts the series negative impedance such that theinput impedance Zin of Port B appears as a shunt negative impedance.

FIG. 1B discloses an embodiment similar to that of FIG. 1A where theseries negative impedance -Z1 has been replaced by a shunt negativeimpedance -Z2. The impedance conversion means 33 controls the inputimpedance of Port B such that the reciprocal (a series negativeimpedance) impedance is exhibited there.

FIGS. 1C and 1D illustrate the impedance conversion means 33 isbidirectional for transmissions. A series negative impedance -Z3connected to Port B will produce a shunt negative input impedance forPort A in FIG. 1C. A shunt negative impedance -Z4 connected to Port Bwill produce a series negative input impedance for Port A in FIG. 1D.

A series or shunt negative impedance can be implemented in numerousways. For example, impedances -Z1, -Z3 could be implemented as a seriesnegative impedance repeater and impedances -Z2, -Z4 as a shunt negativeimpedance repeater. It is further evident that combinations ofimpedances (series or shunt), (negative or positive) can be implementedat either port, either alone or in concert with other impedancecombinations at the opposite port. For example, the implementationsillustrated in FIGS. 1A-1D will in general have a positive loadimpedance in series or parallel with the negative impedances shown.

It is further contemplated by the invention that gain or negativeimpedance can be provided by the impedance conversion means 32 and 34while they additionally provide reciprocal impedance conversion. Withreference now to FIG. 2 there is illustrated another embodiment of theinvention for providing reciprocal negative impedance. The network issimilar in operation and structure to that of FIG. 1 with the additionof means 36 for generating a negative feedback voltage Vf from controlvoltage Vab. The feedback voltage Vf is a selected fraction K of thevoltage Vab which is then inverted in polarity before being added to thevoltage -I₂ Zb to become the control voltage Vba at the secondaryterminal 24 of transformer 16. In operation, when transmitting from PortA to Port B, the negative feedback voltage Vf reduces the voltage Vba (apositive impedance). The result is an increase in I₁ and the controlvoltage Vab, which is the product I₁ ×Za, thereby producing a signalgain by driving the load impedance Zlsb. The control voltage Vba,therefore, appears as a series negative impedance at Port A. Inreversing transmission direction from Port B to Port A, the feedbackvoltage Vf increases Vba (I₁ is reversed) and hence current -I₁ therebyproducing a signal gain by driving the load impedance Zlsa. Again theincrease in -I₁ increases Vab which increases the positive impedance toPort B. This has the effect of reducing I₂ relative to the voltageapplied by the signal generator at Port B. Because of the signal gainand increase in port impedance, the control voltage Vab, therefore,appears as a shunt negative impedance. It should be especially notedthat the FIG. 1A and FIG. 2 implementations perform the same functions,i.e., a series negative impedance is exhibited at Port A and a shuntnegative impedance at Port B. These implementations are particularlyadapted for use as a reciprocal negative impedance repeater for thenonloaded cable facilities of a telephone system where Port A isconnected to the switching equipment and Port B is connected to thenonloaded cable facilities.

Further, it is evident that other feedbacks can be made from theimpedance conversion means 32, 34 to change the characteristics of thecircuit. For example, if the implementation of FIG. 1C were to be madeusing FIG. 4, then a negative fraction of control voltage Vba would becombined with voltage Vab. Other feedbacks are possible and thefollowing rules apply. If the feedback increases the voltage from thesource driving the load, then gain or negative impedance will be seenreflected to the source port. If the feedback voltage decreases theinput impedance of the port while exhibiting gain, it will cause theport to appear as though a series negative impedance were connected. Ifthe feedback voltage increases the input impedance while exhibitinggain, it will cause the port to appear as though a shunt negativeimpedance were connected.

FIGS. 3A-3D will now be more fully explained to illustrateimplementations of the impedance conversion means 32 and 34. FIG. 3aillustrates the generation of the control voltage Vab by an operationalamplifier 50. The noninverting input of the operational amplifier 50 isgrounded while the control impedance Za is connected between theinverting input and output of the amplifier. The inverting input isfurther connected to the terminal 26 of the secondary of the transformer16 of Port A and the output of the amplifier generates the controlvoltage Vab. In this configuration, the operational amplifier 50measures the current I₁ needed to keep the inverting input at the samereference voltage level as the noninverting input and generates thatcurrent through the impedance Za thereby producing the voltage Vab.

Another circuit implementation is illustrated in FIG. 3B to generate thecontrol voltage Vab. An operational amplifier 52 has its noninvertinginput grounded and its output connected to the terminal 26 of thesecondary of the transformer 16 through the control impedance Za. Afeedback resistor Rf is further connected between the control terminaland the inverting input of the amplifier 52. The control voltage Vab isgenerated from the output of the amplifier. The operational amplifier 52measures the amount of current I₁ which is required to keep thenoninverting terminal of the amplifier at ground. This reference voltagelevel is sensed by the feedback resistor Rf. The necessary current I₁through the impedance Za generates the control voltage Vab.

FIG. 3C is a circuit implementation for generating the control voltageVba. An operational amplifier 54 has its noninverting input grounded anda control impedance Zb connected between its output and inverting input.The inverting input is further connected to the terminal 30 of thesecondary of transformer 18. The output of amplifier 54 is connected toan inverting voltage amplifier 56 having a nominal gain of unity. Thecircuit in FIG. 3C operates similarly to that of FIG. 3A where theoperational amplifier 54 measures the amount of current necessary tomaintain the inverting input of the amplifier 54 at ground therebygenerating a voltage I₂ ×Zb. This control voltage is inverted by theoperational amplifier 56 to become the control voltage Vba.

With respect now to FIG. 3D there is shown another preferredimplementation of the impedance conversion means 34. The circuitincludes an operational amplifier 58 having its noninverting terminalgrounded and connected to the secondary terminal 30 of the transformer18 through the control impedance Zb. A feedback resistor Rf is connectedbetween the terminal 30 and the inverting input of the amplifier. Thevoltage generated at the output of the amplifier 58 is transferred to ainverting voltage amplifier 60 to become the control voltage Vba at thesecondary terminal 24 of transformer 16. The operation of thisconfiguration of impedance conversion means 34 is similar to that shownin FIG. 2b. The current I₂ needed to maintain the inverting input atground, which condition is sensed through resistor Rf, is measured byamplifier 58. The voltage generated, which is I₂ ×Zb, is then invertedin the amplifier 60 to become the control voltage of Vba.

FIG. 4 illustrates a detailed circuit implementation of the blockdiagram of FIG. 2 utilizing the circuit implementations of the impedanceconversion means 32 and 34 illustrated in FIGS. 3B and 3D. In additionto the previous elements, a feedback impedance Zx has been added toreflect a fraction of the control voltage Vab back to the Port A sidethrough the inverting amplifier 60 so as to produce gain.

The equations for such implementation, again assuming idealtransformers, are illustrated below:

When transmitting from Port A to Port B, Esb=0, K=R/Zx ##EQU1## It isevident that equation (3) is the equation for a series negativeimpedance, and equation (6) is the equation for a shunt negativeimpedance.

In FIG. 5 another preferred implementation of a reciprocal negativeimpedance repeater constructed in accordance with the invention isshown. A frequency dependent gain control has been added to enhance thestability and equalization of an implementation of a reciprocal negativeimpedance repeater, such as that illustrated in FIG. 4. The gain controlconsists of two relatively independent parts which vary the gain andimpedance of the repeater to compensate for different lengths and gaugesof nonloaded cable and their frequency dependent loss and impedancechanges. In one part, an incremental gain control is used to match thegain versus frequency slope to cable length and gauge by varying thefeedback to the switching equipment side of the repeater. Another partis used to provide optioning for different cable configurations wherethe overall level of frequency dependent gain is matched to theimpedance changes due to cable gauge. The gain control is in general anfrequency dependent circuit which varies the feedback to the switchingequipment side of the repeater to modify its gain and impedance inaccordance with those changes which it perceives in the cable facilitieswhich are attached to its other side.

The repeater illustrated in FIG. 5 is specifically adapted to beconnected on the Port A side at terminals 100-102 to the switchingequipment of a telephone system. On the Port B side of the repeater atterminals 104 and 106, the repeater is specifically adapted to beconnected to the nonloaded cable facilities of the telephone system. Therepeater is bidirectional and linear for transmission from Port A toPort B or from Port B to Port A.

The repeater includes a transformer 108 with a split primary and aone-to-one transformer ratio on the Port A side and a transformer 110with a split primary and a one-to-one transformer ratio on the Port Bside. The two equal split primary coils 112 and 114 couple ACtransmission signals to the secondary coil 116 of the transformer 108and bypass battery DC and ringing current to the split primary coils 118and 120 of the Port B transformer 110. A blocking capacitor 122 of 2.15microfarads is used to maintain DC separation between the two conductorsand present a predetermined capacitance to the input Port A. AC signalsof the voice frequency spectrum pass through an impedance converter tothe secondary coil 124 of transformer 110 and are output from the dualprimary coils 118 and 120 of the transformer 110. Transmission from PortB to Port A reverses the path but is essentially similar.

The current I₁ in the secondary winding 116 of transformer 108 ismeasured by impedance conversion means 126 which outputs a controlvoltage Vab to the secondary 124 of transformer 110. This signal passesthrough three series resistances 128, 130 and 132 which can beselectively inserted and removed from the path, as will be more fullyexplained hereinafter. On the Port B side the impedance conversionmeans, including voltage source 134 and inverting voltage amplifier 136,measure the current I₂ in the secondary 124 of transformer 110 toproduce the control voltage Vba at the secondary 116 of transformer 108.A portion of the voltage Vab is fed back to the inverting amplifier 136by means of the gain control circuitry 138, 140, 142 and resistor 144 aswill be more fully explained hereinafter. The circuit provides thisconfiguration with reciprocal negative impedance in each transmissiondirection.

Further included is a singing detector 142 which along with shortingjack 144 is used to perform an electronic alignment and unconditionalstability test for the repeater. A conductive shorting plug inserted inshorting jack 144 will connect the switching equipment terminals oftransformer 108 directly together to short circuit Port A. Insertion ofthe shorting plug in jack 144 will also open terminals 100-102 todisconnect the switching equipment. The singing detector 142 isconnected through a decoupling capacitor 226 to the output of impedanceconversion means 126. If the voltage is oscillating above a certainfrequency, indicating that the repeater is singing, the AC voltage willbe passed to the base of a PNP transistor 234 where it will develop avoltage across resistor 228. That voltage will turn the transistor 234on such that it provides a conduction path through an LED 230, a loadresistor 232, and its emitter to collector terminals. The conduction ofcurrent through the path will light the LED 230 to provide a visualindication that the repeater is in an unstable or singing mode.

A control impedance 146 is provided between the secondary 116 of thetransformer 108 and the output of the inverting voltage amplifier 136.The control impedance 146 forms part of a built in program for therepeater to determine its initial operating point and is for stabilitypurposes. A matching impedance including a resistor 148 and a capacitor149 is provided in parallel across the primary of transformer 108 onPort A for out of band compensation.

The impedance conversion means 126 and 134 which are identical circuitswill now be more fully explained as to their functional operation. Forexample, the impedance conversion means 134 includes an operationalamplifier 150 which has its noninverting input grounded and its outputconnected through an impedance Zb to the secondary 124 of transformer110. The complex impedance Zb is made up of the combination of resistors147, 152, 154, 156 and a capacitor 158. The resistors form a T-networkwhere a shunt leg is provided between two series legs on either side.The shunt leg changes value with respect to frequency due to theparallel connection of resistor 156 with capacitor 158. Therefore, arepresentation of the complex impedance Zb as a function of frequencyis: ##EQU2##

By measuring the voltage at the reference terminal of secondary 124 oftransformer 110 through resistor 160, the operational amplifier 150maintains its inverting terminal at ground. It generates a current I₂through Zb thereby producing the inverse of the voltage Vba at itsoutput. This voltage is then inverted in the inverting amplifier 136 tobecome Vba. Similarly, for the impedance conversion means 126, theimpedance Za is connected between the output of an operational amplifier162 and the reference terminal of secondary 116 of transformer 108. Thecomplex impedance Za as a function of frequency is: ##EQU3## The voltageat the reference terminal of secondary 116 of transformer 108 is sensedthrough resistance 180 and input to the inverting terminal of theamplifier 162. The amplifier 162 varies its output to maintain itsinverting terminal at ground. Vab is generated by measuring the currentthrough the impedance Za which is produced to provide I₁.

It is noted by definition that Vab and Vba are applied to the terminalsof the transformers 108, 110 and I₁, I₂ are the currents which arecaused to circulate in the ports. Therefore, Za=Vab/I₁ and Zb=-Vba/ I₂which may include impedance contributions in addition to the Za, Zbshown in the preferred embodiment. This analysis is, of course, withresistor 144 open such that there is no feedback. When resistor 144 isconnected in the circuit as shown, Vba is equal to a Vba' which isreduced by the fraction of the voltage Vab which is fed back throughthat circuit path. This feedback, as noted with respect to FIG. 4 andthe equations pertaining thereto, provides gain and influences thegeneration of Vab via its effect on I₁. In general, Vba' appears to PortA as a positive reciprocal impedance which is reduced in proportion tothe feedback or gain in that direction.

The combination of circuits 140, 142 and resistor 144 provides afrequency dependent feedback for the generation of negative impedance orgain. The circuit 140 includes an operational amplifier 182 having anoutput resistor 184 connected between its output and a node 186. Aresistor 188 is connected in series with a plurality of incremental, butnot necessarily equal, resistances 190. A slide switch 192 is used toselect varying multiples of the resistances 190 by moving the slide 194from the left to the right as shown in the drawing. Movement of slide194 connects increasing increments of resistances 190 in series withresistor 188. The terminal of slide switch 192 is connected to theinverting input of the amplifier 182. The circuit 140 further includesthe parallel combination of a resistor 196 and a capacitor 198 connectedbetween the node 186 and the noninverting input of the amplifier 182.The noninverting input of the amplifier 182 is further referencedthrough a resistor 200.

In this configuration the circuit 140 performs as an active inductorwhereby increasing increments of resistance from slide switch 192increases the impedance between the node 186 and the reference voltage.This provides a frequency dependent adjustment for a parallel RLC tankcircuit of which circuit 140 is the L. The operational amplifier 182acts to balance the voltages appearing at its inverting and noninvertinginputs by controlling the current Is through an equivalent resistance Rsconnected between its output and node 186. The equivalent resistance Rsis the parallel combination of resistor 184 and resistor 188 along withany of the added resistors 190. Adding resistors 190 in series withresistor 188 serves to increase the equivalent resistance Rs anddecrease the current Is for a given voltage between node 186 and theinverting input.

The voltage between these two points is frequency dependent because ofcapacitor 198 and Is will vary not only because of slide switch 192 butalso with frequency. The voltage at the noninverting input (andconsequently at the inverting) is set by the divider voltage provided byresistors 196 and 200. At low frequencies the voltage approximates:##EQU4## where the effects of the capacitor 198 are nominal. When thevoltage at node 186 increases in frequency, the capacitor 198 changesimpedance and increasingly shunts resistor 196. Therefore, as frequencyincreases, the voltage at the noninverting input increases toward V₁₈₆where the capacitor 198 completely shorts the resistor 196. As thevoltage rises with frequency at the noninverting input, it also risescomparably at the inverting input to decrease the voltage differenceacross Rs with a consequent decrease in Is. Decreasing the current Isthrough Rs causes the circuit 140 to appear to node 186 as an impedancewhich increases with frequency, and therefore an inductor.

The circuit 142 is the parallel combination of a resistor 204, acapacitor 202, and a plurality of capacitors 206, 208, 210 and 212 whichcan be switched in parallel with the resistor 204 and capacitor 202 viaoptioning switches 214, 216, 218 and 220 respectively. Together circuit140 and 142 form a parallel RLC tank circuit whose capacitance C can beadjusted by closing different switches 214, 216, 218 or 220 to combinecapacitance with capacitor 202 whose inductance L can be changed byvarying the slide 194 of slide switch 192, and whose R is resistor 204.The RLC circuit is connected in parallel at node 186 with the feedbackresistor 144. The control voltage Vab is coupled to the node through anetwork comprising resistor 222 and capacitor 224.

The parallel tank circuit presents a varying complex impedance in shuntwith resistor 144 to change the fraction of the negative feedbackvoltage applied to Port A. The fraction is R₁₇₈ /Zx where Zx is theshunt combination of resistance R₁₄₄ and Zt, the impedance of theparallel tank. The larger the impedance Zt is, the larger the fractionof the voltage at node 186 which is fed back to Port A.

Thus, when slide switch 192 is used to decrease Is, or when increases infrequency cause a decrease in Is, the impedance of the tank circuitincreases, and with it the gain or fraction of the voltage V₁₈₆ fed backto Port A. The circuit 140 provides a gain with a positive slope whichincreases with frequency to offset the increasing loss with frequencyfor nonloaded cable. The slide 192 varies the slope to match the gaugeand length of cable attached.

The gain circuit is manufactured with the ability to create singing orinstability with the maximum impedance of the cable facilities connectedat Port B. This will provide the basis for a convenient alignment testas discussed hereinafter. If the gain is originally set just under thispoint, then the repeater will be stable for all other conditions becausethe impedance of the cable facilities will decrease with length andfrequency, and the cable port becomes more stable with decreasingimpedance. In general, the maximum impedance of the cable facilitieswill occur at DC, but taking into account the inductance of transformer110, the effective maximum impedance of the illustrated circuit occursabout 270 Hz. This frequency is still considered out of band for the VFRand can be used as a stability test point.

To increase instability for setting the test point where desired, thegain around the repeater loop can be increased by decreasing the valuesof resistors 146, and 196. These values are factory selected so that therepeater will be stable at a selected point with the highest gainsetting of the slide switch 192. In general, this is set with thelongest loop (15 kft.) and smallest gauge cable (26 g) for which therepeater is designed. An adequate gain range for shorter loop lengths isprovided by the other settings of the slide switch 192. In this manner,the gain can be increased to the point just under where instabilityoccurs. Thus, as frequency increases and gain must increase to make upfor increasing loss, it will not cause the repeater to become unstable.This is because the highest gain slope has been set for the worst caseimpedance and frequency and the conditions affecting the stability ofthe repeater only improve thereafter.

The option switches 204, 206 and 208 are used to change the Q of thedetuned tank circuit to increase the feedback level at higherfrequencies. Such addition of capacitance not only increases the gainslope with frequency but also moves the slope toward higher frequenciesto shift the level of the feedback. The resistors 128, 130, and 132 areused with the option capacitors 206, 208, and 210 respectively to reducethe amount of I₂ driving the secondary of transformer 110. With thecoarser gauge cables, for example 24 gauge and especially 22 gauge, Vabcauses a larger amount of I₂ in driving the cable because of their lowerimpedance. However, due to the reciprocal nature of the network, thiscreates positive impedance at Port A which requires more feedback toobtain the gain needed. The resistors 128, 130 and 132 reduce 12,thereby decreasing Vba and enhancing the effect of the feedback. Bysetting the level for the feedback by the optioning capacitors andresistors, the range of the slide switch can supply the needed frequencyvariable gain with the correct slope for different ranges or cablegauges.

When switch 214 is closed capacitor 206 is placed in parallel with theRLC tank circuit and resistor 128 is inserted in the serial path fromVab to terminal. When switch 216 is closed capacitor 208 is inserted inthe tank circuit and resistor 130 is inserted in series in the path fromVab to terminal. When switch 218 is closed capacitor 210 is inserted inthe parallel tank circuit and resistor 132 inserted in series in thepath between Vab and terminal. Capacitor 212 is inserted in the paralleltank circuit by closing switch 220. The switches 214, 216, 218 and 220are used individually to provide settings for different cable types and,if not set, their associated resistances are shorted out and theirassociated capacitors are not in the circuit.

FIG. 6 shows an actual example of how the frequency dependent gaincircuit equalizes the losses in nonloaded cable. Graphicalrepresentation A represents the increasing loss in dB for a 26 gaugenonloaded cable 15 kft in length as a function of frequency over thevoice band. Graphical curve B represents the loss of a repeatered cableof the same type and length using the circuit implementation shown inFIG. 5. It is seen that the loss is essentially flat over the voicefrequency band from 300-3000 Hz.

The electronic alignment of the repeater illustrated in FIG. 5 and atest for unconditional stability are easily accomplished with only onetest person and without specialized equipment. A shorting plug isinserted in shorting jack 144 which connects terminals 100, 102 togetherto provide a "worst case" condition for Port A. The nonloaded cable isleft on hook or idle to provide a "worst case" condition for Port B. Thegain slide switch 194 is then advanced one increment at a time until therepeater sings. The singing detector visually indicates when stabilityis not maintained, and the operator then backs the slide switch 194 offone increment so that the LED 230 does not light. At this point therepeater is unconditionally stable and the shorting plug in jack 144 isremoved.

While the present invention has been illustrated and described inconjunction with the various preferred embodiments, it is to beunderstood that numerous changes and modifications may be made theretowithout departing from the scope of the present invention as ishereinafter defined in the appended claims.

What is claimed is:
 1. A bidirectional telephone signal amplifier andreciprocal impedance convertor for amplifying telephone signals flowingbetween first and second telephone lines, comprising:a first portconnecting to said first telephone line; a second port connecting tosaid second telephone line; impedance conversion means connecting tosaid ports for transferring telephone signals between the portscomprisinga first current to voltage converter including first means forsensing an input current and generating an output voltage proportionalto said input current connected to the first port to receive and tosense a first telephone signal input current flowing through the firstport and to and to produce a first telephone signal output voltage inresponse to the first telephone signal input current, the firsttelephone signal output voltage being applied to the second port as asecond port telephone signal output voltage, and a second current tovoltage converter including second means for sensing an input currentand generating an output voltage proportional to said input currentconnected to the second port to receive and to sense a second telephonesignal input current flowing through the second port and to produce asecond telephone signal output voltage in response to the secondtelephone signal input current, the second telephone signal outputvoltage being applied to the first port as a first port telephone signaloutput voltage; means associated with said impedance conversion meansfor providing amplification of telephone signals passing between saidfirst and second ports; and wherein each port has two terminals, andwherein across the terminals of each port is connected a serial circuitincluding low impedance means for sensing the port's incoming telephonesignal current connected in series with means for generating the port'soutgoing telephone signal voltage, said means for sensing being part ofone of said first and second current to voltage convertors, and saidmeans for generating being part of the other of said first and secondcurrent to voltage convertors.
 2. A bidirectional telephone signalamplifier in accordance with claim 1 wherein said second current sensingand voltage generating means includes inversion means for inverting itsgenerating output voltage's polarity.
 3. A bidirectional telephonesignal amplifier in accordance with claim 2 wherein the means forproviding amplification comprises a telephone signal path feeding thegenerated output voltage of said first current sensing and voltagegenerating means to said second current sensing and voltage generatingmeans such that the inverted voltage generated by said second currentsensing and voltage generating means has added to it a componentproportional to the current sensed by said first current sensing andvoltage generating means.
 4. A bidirectional telephone amplifier inaccordance with claim 1 wherein said means for sensing comprises anoperational amplifier having its inverting input connected to a firstinput terminal of a port, having its output connected through aresistive impedance to said first input terminal of a port, and havingits noninverting input connected to the second input terminal of saidport by the means for generating the port's outgoing telephone signalvoltage.
 5. A bidirectional telephone signal amplifier in accordancewith claim 4 wherein said second current sensing and voltage generatingmeans includes inversion means for inverting its generated outputvoltage's polarity.
 6. A bidirectional telephone signal amplifier inaccordance with claim 5 wherein the means for providing amplificationcomprises a telephone signal path feeding the generated output voltageof said first current sensing and voltage generating means to saidsecond current sensing and voltage generating means such that theinverted voltage generated by said second current sensing and voltagegenerating means has added to it a component proportional to the currentsensed by said first current sensing and voltage generating means.